Method for reducing IM2 noise in a down conversion circuit

ABSTRACT

The present invention relates generally to communications, and more specifically to a method and apparatus for minimizing DC offset and second-order modulation products (IM2 noise) while demodulating RF signals. The principle of the invention can be applied to differential, down-conversion circuits ( 50 ) consisting of two differential mixers ( 54, 56 ) in series, a follows: a pair of current sources Ia and Ib are used to provide current to positive and negative channels of the first differential mixer ( 54 ). Providing current to the amplifying transistors of the first mixer ( 54 ) reduces the current drawn through the active mixer switches, reducing the noise generated. The current sources  1   a  and  1   b  are trimmed in a complementary manner where  1   a= I+Delta 1 , and  1   b =Delta 1 . The value of Δ 1  can be determined in a number of manners; for example, it could be established by testing after the circuit has been fabricated, and the value stored on-chip, for future use.

The present invention relates generally to communications, and morespecifically to a method and apparatus for minimizing DC offset andsecond-order modulation products (IM2 noise) while demodulating RF(radio frequency) signals. The preferred embodiment of the inventionmarks a significant advance in satisfying the need for an inexpensive,high-performance, fully-integrable, receiver or transceiver.

BACKGROUND OF THE INVENTION

Many communication systems modulate electromagnetic signals frombaseband to higher frequencies for transmission, and subsequentlydemodulate those high frequencies back to their original frequency bandat the receiver. The original (or baseband) signal may contain, forexample: data, voice or video content These baseband signals may beproduced by transducers such as microphones or video cameras, becomputer generated, or be transferred from an electronic storage device.In general, the use of high frequencies provides longer range and highercapacity channels than baseband signals, and because high frequencysignals can effectively propagate through the air, they can be used forwireless transmissions as well as hard-wired or wave-guidedcommunications.

All of these signals are generally referred to as RF (radio frequency)signals, which are electromagnetic signals; that is, waveforms withelectrical and magnetic properties within the electromagnetic spectrumnormally associated with radio wave propagation.

Wired communication systems which employ such modulation anddemodulation techniques include computer communication systems such aslocal area networks (LANs), point-to-point communications, and wide areanetworks (WANs) such as the internet. These networks generallycommunicate data signals over electrically conductive or optical fibrechannels. Wireless communication systems which may employ modulation anddemodulation include those for public broadcasting such as AM and FMradio, and UHF and VHF television. Private wireless communicationsystems may include cellular telephone networks, personal pagingdevices, HF radio systems used by taxi services, microwave backbonenetworks, interconnected appliances under the Bluetooth standard, andsatellite communications. Other wired and wireless systems which use RFmodulation and demodulation would be known to those skilled in the art.

The focus of this document is on down-conversion or demodulation; theconversion of high frequency signals to lower frequency levels. In thecase of a wireless RF receiver, for example, demodulation wouldtypically consist of down-converting a received signal from its carrierfrequency to baseband.

Most RF receivers use the “super-heterodyne” topology fordown-conversion, which provides good performance in a limited scope ofapplications, such as in public-broadcast FM radio receivers. As will beexplained, the super-heterodyne's limitations make its use in moresophisticated modern applications expensive, and its performance poor.

The super-heterodyne receiver uses a two-step frequency translationmethod to convert an RF signal to a baseband signal. FIG. 1 presents ablock diagram of a typical super-heterodyne receiver 10. The mixerslabelled M1 12 and M2 14 perform the task of translating the RF signalto baseband, while the balance of the components amplify the signalbeing processed and filter noise from it.

The RF band pass filter (BPF1) 18 first filters the signal coming fromthe antenna 20 (note that this band pass filter 18 may also be aduplexer). A low noise amplifier 22 then amplifies the filtered antennasignal, increasing the strength of the RF signal and reducing the noisefigure of the receiver 10. The signal is next filtered by another bandpass filter (BPF2) 24 usually identified as an image rejection filter.The signal then enters mixer M1 12 which multiplies the signal from theimage rejection filter 24 with a periodic signal generated by the localoscillator (LO1) 26. The mixer M1 12 receives the signal from the imagerejection filter 24 and translates it to a lower frequency, known as thefirst intermediate frequency (IF1).

Generally, a mixer (such as M1 12 or M2 14) is a circuit or device thataccepts as its input two different frequencies and presents at itsoutput:

-   (a) a signal equal in frequency to the sum of the frequencies of the    input signals;-   (b) a signal equal in frequency to the difference between the    frequencies of the input signals; and-   (c) the original input frequencies.    The typical embodiment of a mixer is a digital switch which may    generate significantly more tones than those stated above.

The IF1 signal is next filtered by a band pass filter (BPF3) 28typically called the channel filter, which is centred around the IF1frequency, thus filtering out the unwanted products of the first mixingprocesses; signals (a) and (c) above. This is necessary to prevent thesesignals from interfering with the desired signal when the second mixingprocess is performed.

The signal is then amplified by an intermediate frequency amplifier(IFA) 30, and is mixed with a second local oscillator signal using mixerM2 14 and local oscillator (LO2) 32. The second local oscillator LO2 32generates a periodic signal which is typically tuned to the IF1frequency. Thus, the signal coming from the output of M2 14 is now atbaseband, that is, the frequency at which the signal was originallygenerated. Noise is now filtered from the desired signal using the lowpass filter LPF 38, and the signal is passed on to some manner ofpresentation, processing or recording device. In the case of a radioreceiver, this might be an audio amplifier and speaker, while in thecase of a computer modem this may be an analogue-to-digital convertor.

Note that the same process can be used to modulate or demodulate anyelectrical signal from one frequency to another.

The main problems with the super-heterodyne design are:

-   -   it requires expensive off-chip components, particularly band        pass filters 18, 24, 28, and low pass filter 38;    -   the off-chip components require design trade-offs that increase        power consumption and reduce system gain;    -   image rejection is limited by the off-chip components, not by        the target integration technology;    -   isolation from digital noise can be a problem; and    -   it is not fully integratable.

The band pass and low pass filters 18, 24, 28 and 38 used insuper-heterodyne systems must be high quality devices, so electronicallytunable filters cannot be used. As well, the only way to use thesuper-heterodyne system in a multi-standard/multi-frequency applicationis to use a separate set of off-chip filters for each frequency band.

Direct-conversion topologies attempt to perform down-conversion in asingle step, using one mixer and one local oscillator. In the case ofdown-conversion to baseband, this requires a local oscillator (LO) witha frequency equal to the carrier frequency of the input RF signal.

However, this technique will generate DC noise signals which interferewith low-frequency information contained in the demodulated basebandsignal. These DC noise signals are particularly difficult to overcomebecause they are typically unpredictable and time-varying. Severalmechanisms which may generate such DC noise signals in direct-conversiontopologies include the following:

-   -   1. local oscillator leakage. Local oscillator (LO) power leaking        to the RF input will result in DC levels at the mixer output        because it will be mixed with itself. Because one of the output        signals from a mixer is the difference between the two        frequencies being mixed together, and the LO is generating a        powerful signal at the same frequency as the carrier frequency        of the incoming signal being demodulated, the LO signal itself        is demodulated to generate a DC signal at the mixer output;    -   2. leakage of channel interferers. DC levels may be created at        the mixer output when large nearby RF signals leak into the        local oscillator port of the mixer and are self-mixed down to        DC;    -   3. offsets due to mismatching in devices on a fully-integrated        implementation;    -   4. 1/f noise at baseband. 1/f noise is noise with a power        spectra that is inversely proportional to the frequency—in other        words, the power of the noise signal is greater close to DC        (direct current). 1/f noise, or “flicker noise” is generated        largely by the charge trapping and de-trapping properties of        MOSFETs; and    -   5. intermodulation products. Mixing generates sum and difference        products from primary signals. Intermodulation products are        distortions of those products, which may be generated by        non-linearities in electronic components, or harmonics in the        signals being mixed.        Hence, there is a potential for large, time-varying DC signals        to interfere with the comparatively low-amplitude signals of        interest, at or near DC, at the output of the demodulator.

A number of attempts have been made to reduce or compensate for thelevel of these DC noise signals, but none have been very effective orpractical:

-   1. Capacitive Coupling    -   Placing a capacitor in series with the signal path will block DC        noise signals but will also block components of the desired        signal near zero frequency.    -   The severity of the data loss is dependent upon the transmission        modulation and signal coding.    -   Capacitive coupling also has the disadvantage that the size of        the capacitors are generally too large for a fully integrated        receiver.-   2. Adaptive Feedback    -   DC noise signals may also be removed by the use of adaptive        feedback that time-averages the suspected DC offset value and        subtracts the corresponding amount from a convenient point along        the receive path. While feedback-based DC-offset reduction        techniques are more effective than capacitive coupling and are        more easily applied to integrated solutions, the following must        be considered when they are applied:    -   a. the increased level of complexity they add to the design;    -   b. since the DC offsets and near DC offsets may be        indistinguishable from the desired data, some amount of training        time is normally required on a periodic basis to determine the        DC offset accurately; and    -   c. if a long-term average of the DC offset is used to estimate        how much offset must be subtracted from the input, then this        technique will not respond well to rapid variations in the DC        offset level; and-   3. Good Matching of Devices    -   Mis-matching of transistors causes noise and adversely affects        performance.    -   The degree of mis-matching increases as component sizes        decrease, so performance and yields drop with highly integrated        applications. Typically, this problem is addressed by using        large device sizes and/or using multiple components in parallel.        Neither of these methods are highly effective and of course,        result in larger components.        Thus, none of the currently used techniques for addressing the        DC noise problem in direct-conversion architectures is        particularly effective.

It is also of note that the continuing desire to implement low-cost,power efficient receivers has led to intensive research into the use ofhighly integrated designs, an increasingly important aspect for portablesystems, including cellular telephone handsets. This has provenespecially challenging as the frequencies of interest in the wirelesstelecommunications industry (especially low-powercellular/micro-cellular voice/data personal communications systems) haverisen above those used previously (approximately 900 MHz) into thespectrum above 1 GHz.

Thus, there is a need for a method and apparatus for demodulation whichaddresses the problems above. It is desirable that this design befully-integratable, inexpensive and high performance. As well, it isdesirable that this design be easily applied tomulti-standard/multi-frequency applications.

SUMMARY OF THE INVENTION

It is therefore an object of the invention to provide a novel method andsystem of modulation and demodulation which obviates or mitigates atleast one of the disadvantages of the prior art.

One aspect of the invention is defined as a circuit for down-convertinga differential input signal x(t) comprising: a differentialtransconductance input cell consisting of separate positive and negativechannels for receiving positive and negative channels of the inputsignal x(t) and amplifying the positive and negative channels of theinput signal x(t); a first differential mixer for receiving theamplified input signal x(t), and mixing the input signal x(t) with afirst mixing signal φ1, to generate an output signal φ1 x(t); a seconddifferential mixer for receiving the signal φ1 x(t) as an input, andmixing the signal φ1 x(t) with a second mixing signal φ2, to generate anoutput signal φ1 φ2 x(t); a pair of current sources Ia and Ib forproviding current to respective outputs of the positive and negativechannels of the differential transconductance input cell, to reduce thecurrent drawn from the first differential mixer, the current sources Iaand Ib being trimmed in a complementary manner where Ia=I+ΔI, andIb=I−ΔI.

Another aspect of the invention is defined as a method of A method ofsignal demodulation for a circuit having a differential transconductanceinput cell consisting of separate positive and negative channels forreceiving positive and negative channels of the input signal x(t) andamplifying the positive and negative channels of the input signal x(t);a first differential mixer for receiving the amplified input signalx(t), and mixing the input signal x(t) with a first mixing signal φ1, togenerate an output signal φ1 x(t); a second differential mixer forreceiving the signal φ1 x(t) as an input, and mixing the signal φ1 x(t)with a second mixing signal φ2, to generate an output signal φ1 φ2 x(t);a pair of current sources Ia and Ib for providing current to respectiveones of the positive and negative channels of the differentialtransconductance input cell, to reduce the drawn from the firstdifferential mixer; the current sources Ia and Ib being trimmed in acomplementary manner where Ia=I+ΔI, and Ib=I−ΔI; the method comprisingthe steps of: injecting a two-tone signal at the input; measuring IM2 atthe baseband output of the circuit; determining the level of ΔI whichminimizes IM2; recording the level of ΔI which minimizes IM2; and usingthe recorded level of ΔI during normal operation of the down-convertor.

A further aspect of the invention is defined as a method ofdown-converting a differential input signal x(t) comprising the stepsof: amplifying positive and negative channels of the input signal x(t)using a differential transconductance input cell consisting of separatepositive and negative channels; mixing the amplified input signal x(t)with a first mixing signal φ1, to generate an output signal φ1 x(t),using a first differential mixer; mixing the signal φ1 x(t) with asecond mixing signal φ2, to generate an output signal φ1 φ2 x(t), usinga second differential mixer, and providing current to respective ones ofthe positive and negative channels of the differential transconductanceinput cell, using a pair of current sources Ia and Ib, reducing thecurrent drawn from the first differential mixer; and trimming thecurrent sources Ia and Ib in a complementary manner where Ia=I+ΔI, andIb=I−ΔI; wherein ΔI can be manipulated to reduce the IM2 and DC offsetin the output signal φ1 φ2 x(t), and wherein matching parameters for themixers can be relaxed.

BRIEF DESCRIPTION OF THE DRAWINGS

These and other features of the invention will become more apparent fromthe following description in which reference is made to the appendeddrawings in which:

FIG. 1 presents a block diagram of a super-heterodyne receiver topologyas known in the art;

FIG. 2 presents a block diagram of a demodulator topology in a broadembodiment of the invention;

FIG. 3 presents a timing diagram showing the development of a pair ofvirtual local oscillator (VLO) mixing signals;

FIG. 4 presents a timing diagram of a set of differential VLO mixingsignals plotted in amplitude versus time, in an embodiment of theinvention;

FIG. 5 presents an electrical schematic diagram of a differentialdemodulator topology in CMOS, in an embodiment of the invention;

FIG. 6 presents a block diagram of a differential active mixer in anembodiment of the invention;

FIG. 7 presents an electrical schematic diagram of an adjustable currentsource in an embodiment of the invention;

FIG. 8 presents a flow chart of a method of determining the trimmingcurrent, ΔI, in an embodiment of the invention;

FIG. 9 presents a diagram demonstrating how order modulation noise (IM2noise) changes with the trimming current, ΔI; and

FIG. 10 presents an electrical schematic diagram of a differentialdemodulator topology in BiCMOS, in an embodiment of the invention.

DETAILED DESCRIPTION OF THE INVENTION

A circuit which addresses a number of the objects outlined above ispresented as a block diagram in FIG. 2. This figure presents ademodulator or down-conversion topology 50 in which a differential inputsignal x(t) is down-converted by mixing it with two mixing signals φ1and φ2. A “differential” signal is simply a signal which is available inthe form of positive and negative potentials with respect to ground.Thus, this circuit handles an input signal in the form of a differentialvoltage, x(t)+, x(t)− as the radio frequency (RF) input signal.

The use of a differential architecture results in a stronger outputsignal that is more immune to common mode noise than a single endedarchitecture such as that of FIG. 1. If, for example, environmentalnoise imposes a noise signal onto the input of FIG. 1, then this noisesignal will propagate through the circuit. If the same environment noiseis imposed equally on the x(t)+ and x(t)− inputs of the differentialcircuit of FIG. 2, then the net effect will be null.

The circuit of FIG. 2 includes a differential transconductance inputcell 52 which has separate positive and negative channels for receivingpositive and negative channels of the input signal x(t). It amplifiesthe positive and negative channels of the input signal x(t), and passesthe amplified signal to the first differential mixer 54. The inputsignal x(t) may come from any source, thus the differentialtransconductance input cell 52 may be connected to various low noiseamplifiers, filters, antennas or other front end components. Typically,the transconductance input cell 52 will have high impedance inputs, butthe invention is not restricted to such an implementation.

The first differential mixer 54 receives the amplified input signal andmixes it with the first mixing signal φ1, generating an output signal φ1x(t). Similarly, the second differential mixer 56 receives the signal φ1x(t) as an input, and mixes it with the second mixing signal φ2, togenerate an output signal φ1 φ2 x(t).

The two mixing signals φ1 and φ2 may be of any form known in the art,including those of the form used in standard super-heterodynearchitectures. In the preferred embodiment described hereinafter, aparticularly useful paradigm of mixing signals referred to as “virtuallocal oscillator” or VLO signals, will be described, but the inventionis not limited to VLO mixing signals.

Any differential mixers known in the art, or pairs of non-differentialmixers could be used for the two mixers in this circuit. For example,the differential mixer described in the co-pending patent applicationfiled in the United States on Mar. 8, 2002, under application Ser. No.10/096,118, and titled “Integrated Circuit Adjustable RF Mixer”, couldbe used.

The particular design parameters for the two mixers 54 and 56 would bedear to one skilled in the art, having the typical properties of anassociated noise figure, linearity response, conversion loss, conversioncompression, isolation, dynamic range, distortion conversion gain. Theselection and design of these mixers would follow the standards known inthe art.

The circuit also includes a pair of current sources Ia and Ib forproviding current to respective outputs of the positive and negativechannels of the differential transconductance input cell 52. Thesecurrent sources reduce the current which must be drawn through the firstdifferential mixer 54.

The operation of the invention will become more clear from the detaileddescription of the preferred embodiment which follows, with respect toFIGS. 5 through 9. In short, the components of the first differentialmixer 54 must draw electrical current in the course of their operationfor amplification and switching purposes (and possibly other functions).Providing current to RF amplifier transistors in the first differentialmixer 54 from an external source such as Ia and Ib, means that theactive mixer switches in the first differential mixer 54 are onlyrequired to provide a small proportion of the current required by the RFamplifier transistors. The reduced contribution of current via theactive mixer switches, results in less noise being referred to theinputs of the RF amplifier transistors and also less noise beinggenerated by the resistive load of the active mixer circuit, resultingin improved overall noise performance. At the same time, the totalcurrent flowing through the RF amplifier transistors can be maintainedat a level sufficient to ensure their operation at the required gain andlinearity.

It is also important that the current sources Ia and Ib be trimmed in acomplementary manner where Ia=I+ΔI, and Ib=I−ΔI. The value of ΔI can bedetermined in a number of ways, including, for example, various feedbackor testing techniques. In the preferred embodiment describedhereinafter, the value of ΔI is determined by performing a two-tone testprior to regular operation of the circuit. The value of ΔI is adjustedduring the course of this test to minimize second order modulationproducts at the output. The optimal value of ΔI is stored and used inthe course of regular operation of the circuit.

As noted in the Background, DC noise terms or DC offsets are generatedin the course of down-conversion. The circuit of the invention providesfor DC offset correction in such away that the matching of devices isrelaxed, making the circuit more robust and providing higher yields inan integrated environment

Many additions and changes may be made to this circuit, and still allowthe concept of the invention to be exploited. For example, the circuitof FIG. 2 may be provided with a filter 58 between the first and secondmixers 54, 56 depending on the nature of the down-conversion model. Inthe case of traditional super-heterodyne conversion, for example, abandpass filter may be used. In the case of VLO conversion describedhereinafter, a high pass filter (HPF) may be used.

Though FIG. 2 implies that various elements are implemented in analogueform, they can also be implemented in digital form. The mixing signalsare typically presented herein in terms of binary 1s and 0s, however,bipolar waveforms, ±1, may also be used. Bipolar waveforms are typicallyused in spread spectrum applications because they use commutating mixerswhich periodically invert their inputs in step with a local controlsignal (this inverting process is distinct from mixing a signal with alocal oscillator directly).

A number of other embodiments of the invention will now be described,but first, the concept of virtual local oscillators (VLOs) will bedescribed.

Virtual Local Oscillator (VLO) Signals

As noted above, it is preferable that the invention be implemented usingvirtual local oscillator (VLO) signals. VLO mixing signals are verydifferent from mixing signals used in normal two-step conversiontopologies (such as super-heterodyne topologies). Though two mixingsignals are used in a VLO implementation, the two VLO signals are morecomparable to the single mixing signal used in direction-conversion. Themain difference from the direct-conversion approach is that two VLOmixing signals are used to emulate the single mixing signal, without theusual short comings of direct-conversion, such as self-mixing. This isbecause the two VLO mixing signals never really generate the LO signalbeing emulated.

When demodulating an input signal x(t) to baseband usingdirection-conversion, the usual practice is to mix the input signal x(t)with a signal f1 at the carrier frequency of the input signal x(t). TheVLO philosophy is to emulate this demodulation using two (or more)mixing signals with a number of special properties:

-   -   1. their product emulates a local oscillator (LO) signal that        has significant power at the frequency necessary to translate        the input signal x(t) to the desired output frequency. For        example, to translate the input signal x(t) to baseband,        φ1(t)*φ2(t) must have significant power at the carrier frequency        f1, of x(t); and    -   2. one of either φ1 and φ2, has minimal power around the        frequency of the mixer pair output φ1(t)*φ2(t)*x(t), while the        other has minimal power around the centre frequency, f_(RF), of        the input signal x(t). “Minimal power” means that the power        should be low enough that it does not seriously degrade the        performance of the RF chain in the context of the particular        application.    -   For example, if the mixer pair is demodulating the input signal        x(t) to baseband, it is preferable that one of either φ1 and φ2        has minimal power around DC.        As a result, the desired demodulation is affected, but there is        little or no LO signal at the carrier frequency of the input        signal x(t), to leak into the signal path and appear at the        output.

FIG. 3 presents an example of a suitable φ1 and φ2 mixing signal pair,which emulates the LO signal f1. In this embodiment, the first mixingoperation (with the first mixer 52 of FIG. 2) is performed with themulti-tonal mixing signal φ1. Multi-tonal, or non-mono-tonal, refers toa signal having more than one fundamental frequency tone. Mono-tonalsignals have one fundamental frequency tone and may have other tonesthat are harmonically related to the fundamental tone.

The resulting signal, φ1 x(t), is then mixed with the mono-tonal signalφ2 by means of the second mixer 54, generating an output signal φ1 φ2x(t).

Looking at FIG. 3, it is clear that the product of these two mixingsignals, φ1*φ2, has significant power at the frequency of the localoscillator signal f1 being emulated. However, neither φ1 nor φ2 havesignificant power at the frequency of the input signal x(t), the f1 LOsignal being emulated, or the output signal φ1 φ2 x(t). Mixing signalswith such characteristics greatly resolves the problem of self-mixingbecause the VLO signals simply do not have significant power atfrequencies that will interfere with the output signal.

It is also important to note that at no point in the operation of thecircuit is an actual “φ1*φ2” signal ever generated and if it is, only aninsignificant amount is generated. The mixers 54, 56 receive separate φ1and φ2 signals, and mix them with the input signal x(t) using differentphysical components. Hence, there is no LO signal which may leak intothe circuit.

Looking at one cycle of these mixing signals from FIG. 3 the generationof the φ1*φ2 signal is clear:

f1 φ1 φ2 φ1 * φ2 LO LO LO LO HI HI LO HI LO LO LO LO HI HI LO HI LO HIHI LO HI LO HI HI LO HI HI LO HI LO HI HI

Clearly, the two mixing signals φ1 and φ2 in FIG. 3 satisfy the criteriafor effective VLO signals.

The design of circuits for the generation of such signals would be clearto one skilled in the art from the teachings herein. A large number ofsuitable circuits are also described in the Applicant's related,co-pending patent applications:

-   1. PCT International Application Serial No. PCT/CA00/00995 Filed    Sep. 1, 2000, titled: “Improved Method And Apparatus For    Up-Conversion Of Radio Frequency (RF) Signals”;-   2. PCT International Application Serial No. PCT/CA00/00994 Filed    Sep. 1, 2000, titled: “Improved Method And Apparatus For    Down-Conversion Of Radio Frequency (RF) Signals”; and-   3. PCT International Application Serial No. PCT/CA00/00996 Filed    Sep. 1, 2000, titled: “Improved Method And Apparatus For    Up-And-Down-Conversion Of Radio Frequency (RF) Signals”.    The problems of image-rejection, LO leakage and 1/f noise in highly    integrated transceivers can be largely overcome by using these VLO    signals.

It would be clear to one skilled in the art that VLO signals may bedesigned which provide the benefits of the invention to greater orlesser degrees. While it is possible in certain circumstances to havealmost no LO leakage, it may be acceptable in other circumstances toincorporate VLO signals which still allow a degree of LO leakage.

Voltage controlled oscillators (VCOs) are typically used to generate VLOmixing signals. As a general rule, it is desirable to use oscillatorswhich operate at frequencies which will not adversely affect the datasignal if any self-mixing occurs, for example using a VCO at a multipleor divisor of the LO signal being emulated.

FIG. 4 presents a timing diagram similar to that of FIG. 3, for thegeneration of differential VLO mixing signals. From the description ofFIG. 3 above, the development of FIG. 4 follows logically.

The goal in this case, is to generate a set of differential mixingsignals φ1P, φ1N, φ2P and φ2N, where φ1P and φ2P combine to emulate thepositive channel of the LO signal (f1P), and φ1N and φ2N combine toemulate the negative channel of the LO signal (f1N). The positive andnegative pairings of VLO signals are simply polar complements of oneanother.

As noted above, the input signal x(t) is down-converted to basebandusing the two mixers 54, 56 and differential mixing signals φ1P, φ1N,φ2P and φ2N. Because differential mixing signals are employed, positiveand negative pairings must be generated for each of φ1 and φ2. Eachpairing of positive and negative signal components are simplycomplements of one another, so the pattern of these signals followslogically from the amplitude versus time graph of FIG. 3. Forcompleteness however, the development of these signals are shown in theamplitude versus time graph of FIG. 4.

In operation, the monotonal signal φ1P is mixed with the x(t)+ input,and then φ1P*x(t)+ is mixed with the non-monotonal φ2P. Clearly, theproduct φ1P*φ2P is equal to f1P, so it emulates the f1P signal withoutgenerating significant power at the f1P frequency. Similarly, inoperation, the monotonal signal φ1N is mixed with the x(t)− input, andthen φ1N*x(t)− is mixed with the non-monotonal φ2N. Again, the productφ1N*φ2N is equal to f1N, so it emulates the f1N signal withoutgenerating significant power at the f1N frequency.

Exemplary Circuit

An exemplary implementation of the invention is presented in theschematic diagrams of FIGS. 5 through 9. FIG. 5 presents an electricalschematic diagram of the complete circuit, while FIG. 6 presents ageneralization of the active mixer component of the topology. FIG. 7presents a detail electrical schematic diagram of the current sourcesused in this embodiment. FIG. 8 presents a flow chart of a methodologyfor determining an optimal value for ΔI, while FIG. 9 presents a diagramshowing how this methodology minimizes the IM2.

The circuit 68 presented in FIG. 5 can be described as having four majorcomponents: a pair of current sources 70, an active mixer circuit 72, ahigh pass filter (HPF) 74, and a passive mixer circuit 76. This circuit68 receives a differential RF input signal x(t)+, x(t)−, anddown-converts this signal to a differential baseband (BB) signal, BB+,BB−. At the LO ports of the two mixers 72, 76 differential mixingsignals φ1P, φ1N, φ2P and φ2N are applied to down-convert the incomingRF signal to baseband. These differential mixing signals φ1P, φ1N, φ2Pand φ2N could be standard super-heterodyne mixing signals, or could beVLO signals as described above.

The core of the topology in FIG. 5 consists of two mixers: a first mixer72 which is active, and a second mixer 76, which is passive. Activemixers are distinct from passive mixers in a number of ways:

-   1. they provide conversion gain. Thus, an active mixer can replace    the combination of a low noise amplifier and a passive mixer;-   2. active mixers provide better isolation between the input and    output ports because of the impedance of the active components; and-   3. active mixers allow a lower powered mixing signal to be used,    reducing the noise that results when the mixing signal is generated.

In spite of these advantages, the application of active mixers inmodulation and demodulation topologies is still problematic. Becauseactive mixers are non-linear devices, they generate more 1/f noise andproduce second-order distortion. As noted above, 1/f noise is noise witha power spectra that increases as the frequency approaches DC (directcurrent).

The topology of the invention can exploit the advantages of an activemixer mainly because of the system for reducing DC offsets, but alsobecause the high pass filter 74 and passive mixer 76 are used in thebalance of the circuit 68. To begin with, the high pass filter 74 blocksout a great deal of the DC noise. Then, because the second mixer 76 is apassive mixer and it operates at a relatively lower frequency, it doesnot introduce a significant amount of second-order distortion into thesignal. Thus, this topology provides the benefits of active mixing,without introducing second-order distortion into the output signal.

The operation of the pair of current sources 70 and active mixer circuit72 will now be described with respect to the block diagram of FIG. 6.

A simplified representation of the current sources 70 and the activemixer 72 is presented in the block diagram of FIG. 5, where thecomponents are collected into three groups: a Mixer Block 220, a GainBlock 222, and a Current Source Block 224.

Briefly, the Gain Block 222 is a gain-providing stage that consists of anumber of input transistors, shown in FIG. 4 as transistors M5 and M6.These input transistors are fed with the differential input signalsx(t)+ and x(t)−, and their outputs are fed to the Mixer Block 220 asamplified signals. The mixer block 220 consists of transistors M1through M4 as shown in FIG. 5.

The Gain Block 222 is simply a single stage differential amplifier,consisting of two transistors M5 and M6, and two resistors R1 and R2.The degree of amplification is controlled via the voltage of the inputsignal Vb.

The Mixer Block 220 is effected by two separate transistor and resistorpairings, which receive the amplified RF signals from the Gain Block222. The amplified RF signal from the Gain Block 222 is passed to thesources of the transistor switches M1, M2 and M3, M4, and the drains oftransistors switches M1, M2 and M3, M4 are connected to load resistorsR3 and R4. By feeding the gates of the switching transistors switchesM1, M2 and M3, M4 with complementary mixing signals φ1P and φN, that is,φ1P=−(φ1N), a differential output signal is received. The value of theload resistors RI and RI is selected to provide the best bias conditionsfor the mixer transistors.

Additional details regarding the design and implementation of a suitableactive mixer 72 are given in the co-pending patent application filedunder Canadian Patent Application Serial No. 2,375,438, titled:“Improvements to a High Linearity Gilbert I Q Dual Mixer”. Other activemixer designs may also be used, as known in the art, or variations onthe above used.

The conditions of operation, and hence performance, of the Gain Block222 are alterable through the Current Source Block 224 which provides avariable amount of biasing current to the Gain Block 222. The CurrentSource Block 224 provides current to the Gain Block 222 so that thiscurrent is not drawn entirely from the Mixer Block 230.

The reduced contribution of current to the Gain Block 222 via the MixerBlock 230 results in less noise being referred to the inputs of the RFamplifier transistors in the Gain Block 222 and also less noise beinggenerated, resulting in improved overall noise performance. At the sametime, the total current flowing through the RF amplifier transistors inthe Gain Block 222 can be maintained at a level sufficient to ensuretheir operation at the required gain and linearity.

The current sources Ia and Ib are arranged to provide the currentrequired by the RF amplifier transistors M5 and M6 thereby requiring theactive mixer switches M1, M2 and M3, M4 to provide only a smallproportion of the current required for the Gain Block 222. This resultsin improved overall noise performance.

The linearity of the active mixer 72 is also improved by this currentinjection between the input amplifier and the active mixer switches M1,M2 and M3, M4 because the current flowing through the input amplifier(Gain Block 222) can be substantially independent of that flowingthrough the active mixer switches M1, M2 and M3, M4.

Outputs from the switching transistors M1, M2 and M3, M4 in the activemixer 72 are then passed through a pair of high pass filters 74, eachconsisting of a capacitor C1 and C2 and two resistors R5 through R8. Theuse of the resistors in the configuration of voltage dividers acrosspositive and negative voltage sources (V_(DD) being positive and V_(SS)being negative) not only serves to drain the capacitors of the high passfilter, but also sets the common mode voltage for the next mixing stage.

Also, note that the cut-off frequency of the pair of high pass filters74 can be very low (either low with respect to the carrier frequency orclose to DC, depending on the application and expected signals). As aresult it may be considered to function almost entirely in the manner ofa voltage divider. Also, the pair of high pass filters 74 might beeffected in other manners, for example, in the form of an ‘activeresistor’ network.

The outputs of the pair of high pass filters 74 are then passed to theinputs of the respective halves of the differential passive mixer 76,whose other inputs are the mixing signals φ2P and φ2N, which work inconcert with the φ1P and φ1N mixing signals used in the active mixer 72.In FIG. 4, the passive mixer 74 comprises a known design having fourtransistors M7, M8, M9 and M10. Other architectures could also be used.

If VLO mixing signals are being used, this second mixing stage completesthe emulation of the local oscillator mixing, frequency translating theinput x(t)+ and x(t)− signal to the desired output signal φ1N φ2N x(t)−and φ1P φ2P x(t)+. If this circuit is being used to demodulate a signaldown to baseband, as it would in the case of a radio receiver, it maythen be desirable to pass the outputs of the passive mixer 76 through alow pass filter to remove any significant out-of-band signals.

One of the further benefits of this design is the use of simpleresistive elements (R1 and R2) to fix the active mixer 72 biasingvoltages. This assists in the selection of optimal performanceparameters for the passive mixer 76.

The linearity of an active mixer is dependent on the biasing voltage oftransistors. There are at least two sources of non-linearity in theactive mixer 72: the non-linearity of the RF amplifier transistors andthat of the switching transistors. The optimum biasing must be foundthrough simulation or other techniques. The bias voltage applied to eachof the drains of the active mixer switches is thereby selected and fixedto that necessary for optimum linearity during design.

FIG. 7 presents an exemplary circuit for implementing the currentsources 70 of the invention. The two current sources are implementedusing parallel arrays of transistors, controlled by electronic switches.The lowest level of Ia current, for example, will be equivalent to thecurrent provided through transistor Ma. This current level can beincreased using switches Sb through Sx to operate transistors Mb throughMx, connected in parallel to transistor Ma. These switches Sb . . . Sxare driven selectively by the level of the ΔI current. While only threetransistors are shown for this channel, clearly a large number could beused.

Similarly, the Ib channel consists of a corresponding array oftransistors M′b through M′x, connected in parallel to transistor M′a.The Ib channel will have a minimum current level determined by thecurrent through transistor M′a, but this current can be increased usingswitches S′b through S′x (which, like the Ia channel, are driven by thelevel of the ΔI current).

This circuit 70 also includes a common mode feedback circuit (CMFB),which receives as an input, the outputs of the active mixer 72, x′ andy′. The CMFB circuit receives this pair of signals and determines thecommon mode level for the two inputs, ensuring that the outputs x′ andy′ have a fixed common mode voltage.

Note that any CMFB circuit know in the art could be used, including thefollowing:

-   1. a switched capacitor design;-   2. a differential difference amplifier (DDA) design;-   3. a resistor-averaged circuit; or-   4. other designs.    Exemplary Method of Determining Trimming Current ΔI

FIG. 8 presents a flow chart of a methodology for determining andapplying the trimming current (ΔI) value. As noted above, the desire isto determine two complementary current values which feed the activemixer 72: Ia=I+ΔI and Ib=I−ΔI. The ΔI value can be controlled usingdigital or/and analog methods, and is used to reduce the DC term at theoutput due to a large single tone input. In the preferred embodiment,the value of the current ΔI is determined as follows:

First, a two tone signal at frequencies f1 and f2 is injected at thex(t) input of the transconductance cell 52 of the circuit, at step 90.The IM2 tone (i.e. the tone at frequency f1-f2) is then measured at theBB output of the circuit, per step 92.

The power level of the IM2 is then minimized by adjusting ΔI per step94, and continuously measuring changes to IM2 at step 92. Once a minimumvalue for IM2 is determined, control passes to step 94. At step 96, thevalue of ΔI which yields the minimum value for IM2, is stored on chip inany method know in the art

This optimal value for ΔI is then used during regular operation of thecircuit, per step 98. If an RF tone which is AM modulated is injected atthe input of the transconductor, the amount of AM detection signal powerat baseband is now minimized as a function of ΔI.

It is generally only necessary to determine the optimal value of ΔI oncefor any given chip, because its value will be determined by fabricationfactors which do not vary greatly over the life of the chip. Thus, thisprocess can be executed in the factory before delivery of the chip.

FIG. 9 presents a graph of how the second order distortion (IM2) isminimized. The x-axis on this figure represents the level of thetrimming current ΔI input to the current sources, and the y-axisrepresents the corresponding level of IM2 distortion in the output atf1-f2. The circuit of the invention will generate a level of IM2distortion which will vary with the ΔI input to the current sources,following a curve like that of FIG. 9, which will have a lowest pointfor some level of ΔI. The task is simply to determine the level of ΔIwhich generates the lowest IM2 distortion.

The invention can be implement using bipolar technology, CMOStechnology, BiCMOS technology, or another semiconductor technology. FIG.10 presents a circuit diagram of a BiCMOS implementation 120 which iscomparable to that of the FIG. 5 CMOS implementation.

The main differences between the BiCMOS and CMOS implementations are:

-   1. the active mixer 72 is implemented using transistors Q1 through    Q6;-   2. a current sink Is is required for the active mixer 72;-   3. the passive mixer is implemented using transistors M1 through M4;    and-   4. the sense of the modulating signals 1P-, 1N-, 2P-, 2N- are    changes to suit the polarities of the transistors as required.

The invention could also be implemented using other fabricationtechnologies including, but not limited to Silicon/Germanium (SiGe),Germanium (Ge), Gallium Arsenide (GaAs), and Silicon on Sapphire (SOS).

Advantages of the Invention

The invention provides many advantages over other down-convertors knownin the art. To begin with, it offers:

-   1. minimal 1/f noise;-   2. minimal imaging problems;-   3. minimal leakage of a local oscillator (LO) signal into the RF    output band;-   4. removes the necessity of having a second LO as required by    super-heterodyne circuits, and various (often external) filters; and-   5. has a higher level of integration as the components it does    require are easily placed on an integrated circuit. For example, no    large capacitors or sophisticated filters are required.

A high level of integration results in decreased IC (integrated circuit)pin counts, decreased signal power loss, decreased IC powerrequirements, improved SNR (signal to noise ratio), improved NF (noisefactor), and decreased manufacturing costs and complexity.

The design of the invention also makes the production of inexpensivemulti-standard/multi-frequency communications transmitters and receiversa reality.

The benefits of the invention are most apparent when it is implementedwithin a single-chip design, eliminating the extra cost ofinterconnecting semiconductor integrated circuit devices, reducing thephysical space they require and reducing the overall power consumption.Increasing levels of integration have been the driving impetus towardslower cost, higher volume, higher reliability and lower power consumerelectronics since the inception of the integrated circuit. Thisinvention will enable communications devices to follow the sameintegration route that other consumer electronic products have benefitedfrom.

Options and Alternatives

A number of variations can be made to the topology of the inventionincluding the following:

-   1. the invention could be implemented in a multi-band/multi-standard    application.    -   A mixer topology that is suitable as part of a        multi-band/multi-standard receiver, is described in detail in        the co-pending patent application filed under the Patent        Cooperation Treaty under application number PCT/CA02/01316,        filed on Aug. 28, 2002, and titled Improved Apparatus And Method        For Down-conversion.    -   The topology shown and described in this co-pending application        is almost the same as that of FIG. 5. The difference is simply        that it offers the added functionality of receiving more than        one RF input, which can be electronically selected. This is        effected simply by means of electronic switches connected to        various RF inputs, the switches being used to control which RF        signal is to applied to the mixing transistors; or-   2. the invention could be implemented using in-phase and quadrature    signals in many modulation schemes, it is necessary to modulate or    demodulate both in-phase (I) and quadrature (Q) components of the    input signal. In such a case, four modulation functions would have    to be generated: φ1I which is 90 degrees out of phase with φ1Q; and    φ2I which is 90 degrees out of phase with φ2Q. The pairing of    signals φ1I and φ2I must meet the function selection criteria listed    above, as must the signal pairing of φ1Q and φ2Q. Design of    components to generate such signals would be clear to one skilled in    the art from the description herein. As well, additional details on    the generation of such signals are available in the co-pending    patent applications filed under PCT International Application Serial    Nos. PCT/CA00/00994, PCT/CA00/00995 and PCT/CA00/00996.

CONCLUSIONS

It will be apparent to those skilled in the art that the invention canbe extended to cope with more than two or three standards, and to allowfor more biasing conditions than those in the above description.

The electrical circuits of the invention may be described by computersoftware code in a simulation language, or hardware development languageused to fabricate integrated Druids. This computer software code may bestored in a variety of formats on various electronic memory mediaincluding computer diskettes, CD-ROM, Random Access Memory (RAM) andRead Only Memory (ROM). As well, electronic signals representing suchcomputer software code may also be transmitted via a communicationnetwork.

Clearly, such computer software code may also be integrated with thecode of other programs, implemented as a core or subroutine by externalprogram calls, or by other techniques known in the art.

The construction of the necessary logic to generate the mixing signalsof the invention would be clear to one skilled in the art from thedescription herein. Such signals may be generated using conventionalmethods and components including basic logic gates, field programmablegate arrays (FPGAs), programmable array logic (PALs) or gate array logic(GALs). The signals of the invention may also be stored on memorydevices such as read only memories (ROMs), programmable read onlymemories (PROMs), erasable programmable read only memories (EPROMs),electrically erasable programmable read only memories (EEPROMS) or flashmemory, and cycled out as required. The embodiments of the invention mayalso be implemented using processor-type devices such as digital signalprocessors (DSPs), microcontrollers, microprocessors, or similar devicesas known in the art. Such implementations would be clear to one skilledin the art.

The embodiments of the invention may be implemented on various familiesof integrated circuit technologies using digital signal processors(DSPs), microcontrollers, microprocessors, field programmable gatearrays (FPGAs), or discrete components. Such implementations would beclear to one skilled in the art.

The invention may be applied to various communication protocols andformats including: amplitude modulation (AM), frequency modulation (FM),frequency shift keying (FSK), phase shift keying (PSK), cellulartelephone systems including analogue and digital systems such as codedivision multiple access (CDMA), time division multiple access (TDMA)and frequency division multiple access (FDMA).

The invention may be applied to such applications as wired communicationsystems include computer communication systems such as local areanetworks (LANs), point to point signalling, and wide area networks(WANs) such as the Internet, using electrical or optical fibre cablesystems. As well, wireless communication systems may include those forpublic broadcasting such as AM and FM radio, and UHF and VHF television;or those for private communication such as cellular telephones, personalpaging devices; wireless local loops, monitoring of homes by utilitycompanies, cordless telephones including the digital cordless Europeantelecommunication (DECT) standard, mobile radio systems, GSM and AMPScellular telephones, microwave backbone networks, interconnectedappliances under the Bluetooth standard, and satellite communications.

While particular embodiments of the present invention have been shownand described, it is clear that changes and modifications may be made tosuch embodiments without departing from the true scope and spirit of theinvention.

1. A method of signal demodulation for a circuit having a differentialtransconductance input cell consisting of separate positive and negativechannels for receiving positive and negative channels of said inputsignal x(t) and amplifying said positive and negative channels of saidinput signal x(t); a first differential mixer for receiving saidamplified input signal x(t), and mixing said input signal x(t) with afirst mixing signal φ1, to generate an output signal φ1 x(t); a seconddifferential mixer for receiving said signal φ1 x(t) as an input, andmixing said signal φ1 x(t) with a second mixing signal φ2, to generatean output signal φ1 φ2 x(t); a pair of current sources Ia and Ib forproviding current to respective ones of said positive and negativechannels of said differential transconductance input cell, to reduce thecurrent drawn from said first differential mixer; said current sourcesIa and Ib being trimmed in a complementary manner where Ia=I+ΔI, andIb=I−ΔI; said method comprising the steps of: injecting a two-tonesignal at said input; measuring IM2 at the baseband output of saidcircuit; determining the level of ΔI which minimizes IM2; recording thelevel of ΔI which minimizes IM2; and using said recorded level of ΔIduring normal operation of said down-convertor.
 2. A computer readablememory medium for storing software code executable to perform the methodsteps of claim
 1. 3. The method of claim 1, including operating a meansfor manipulating ΔI to reduce the IM2 and DC offset in the output signalφ1 φ2 x(t), whereby matching parameters for said mixers can be relaxed.4. The method of claim 1, including operating a means for setting alevel of ΔI.
 5. The method of claim 1, wherein the current sources Iaand lb each include parallel arrays of transistors, and the step ofusing includes selectively driving the parallel arrays of transistorswith a level of ΔI.
 6. A method of down-converting a differential inputsignal x(t) comprising the steps of: amplifying positive and negativechannels of said input signal x(t) using a differential transconductanceinput cell consisting of separate positive and negative channels; mixingsaid amplified input signal x(t) with a first mixing signal φ1, togenerate an output signal φ1 x(t), using a first differential mixer;mixing said signal φ1 x(t) with a second mixing signal φ2, to generatean output signal φ1 φ2 x(t), using a second differential mixer; andproviding current to respective ones of said positive and negativechannels of said differential transconductance input cell, using a pairof current sources Ia and Ib, reducing the current drawn from said firstdifferential mixer; and trimming said current sources Ia and Ib in acomplementary manner where Ia=I+ΔI, and Ib=I−ΔI; wherein ΔI can bemanipulated to reduce the IM2 and DC offset in the output signal φ1 φ2x(t), and wherein matching parameters for said mixers can be relaxed. 7.The method of claim 6, wherein ΔI is determined using a two-tone test,Al being the current level which minimizes IM2 output at baseband. 8.The method of claim 7, wherein the two-tone test includes injecting atwo-tone signal as the input signal x(t); measuring IM2 of the outputsignal φ1 φ2 x(t); determining the level of ΔI which minimizes IM2;recording the level of ΔI which minimizes IM2; and using said recordedlevel of ΔI during the step of providing current.
 9. The method of claim6, wherein the current sources Ia and Ib each include parallel arrays oftransistors, and the step of providing current includes selectivelydriving the parallel arrays of transistors with a level of ΔI.